A superheterodyne receiver is currently the most common type of receiver used in modern communications devices. Such receivers can be found in virtually any home, office, or automobile within a television set, telephone, or radio. A superheterodyne receiver mixes (or multiplies) an incoming radio-frequency (RF) signal (carrier at frequency f1) with a sinusoid signal (at a frequency f2) generated by a local oscillator. The resulting output signal comprises two frequency-shifted versions of the incoming signal centered at the sum and difference of the combining frequencies (f1+f2 and f1−f2).
Typically, the highest frequency components (centered at f1+f2) are filtered out using a bandpass filter and the output signal only contains the intermediate-frequency (IF) components (centered at f1−f2). This process may be repeated several times in high-performance superheterodyne receivers.
While superheterodyne receivers are widely used, superheterodyne receivers use expensive and non-integrable RF and IF components such as bandpass filters. As a result, superheterodyne receivers are not ideal for applications in small, low cost mobile communication systems such as cellular phones, pagers, cordless phones, and the like.
Alternative receivers, such as a direct conversion receiver, are well known in the art and potentially offer significant advantages over the superheterodyne receiver.
A traditional direct conversion receiver, as shown in FIG. 1, directly converts an incoming signal into its baseband in-phase and quadrature components without any intermediate translation into an IF signal. The operation of this traditional direct conversion receiver is simple.
An incoming bandpass signal g(t) (which can be mathematically represented by g(t)=gi(t)cos(2f1t)−gq(t)sin(2f1t) is received at the RF input and then passed through a preselector filter 1 and a low-noise amplifier (LNA) 2. The preselector filter 1 is simply a bandpass filter designed to pass the desired signal g(t) and to reject spurious out-of-band signals. In most applications, the bandwidth of the preselector filter 1 is much greater than the bandwidth of the desired signal. Furthermore, the preselector filter 1 may pass unwanted signals in addition to the desired signal.
After passing through the preselector filter 1, the signal g(t) is split and sent through the two mixers 3, 3′. In the upper mixer 3′, the signal g(t) is mixed with a sinusoid tuned to the same frequency as the carrier frequency (e.g., cos(2 f1t)). In the lower mixer 3, the signal g(t) is mixed with the same sinusoid as in the upper mixer 3′, but with a phase change of /2 (e.g., sin(2 f1t)). The mixers 3, 3′ produce the in-phase (gi(t)) and quadrature (gq(t)) components of the desired signal (g(t)) centered at baseband and at twice the carrier frequency (2fc). The high frequency components are eliminated by the low pass filters 6, 6′, and the in-phase and quadrature signals are finally amplified by the amplifiers 7,7′. There are several advantages of a direct conversion receiver over the more popular superheterodyne receiver.
First, the direct conversion receiver directly converts the incoming signal into its baseband signal directly and eliminates the step of initially translating the RF signal into an IF signal. Thus, all of the intermediate filters, mixers, and amplifiers can be omitted and the circuit is simplified.
Secondly, exception of the preselector filter, the direct conversion receiver employs only low pass filters rather than bandpass filters. Normally, it is easier to integrate a low pass filter onto a single chip than a band pass filter. Thus, the direct conversion receiver may be largely constructed on a single integrated circuit, which makes it smaller and less expensive than a superheterodyne receiver.
While there are advantages to direct conversion receivers over superheterodyne receivers, the traditional direct conversion receiver suffers from some disadvantages.
One problem with traditional direct conversion receivers is second-order distortion present in the mixer. Second-order distortion is caused by the fact that a mixer is inherently a non-linear device. When an off-channel RF signal is detected along with the desired signal, the non-linearity in the mixers produce the second harmonic of the undesired signal at baseband plus a DC offset. Since the direct conversion receiver also shifts the desired signal to baseband, this second-order distortion produced by the mixer can significantly reduce the performance of the receiver.
Moreover, the mixer can operate like a “square law” detector and convert the envelope of a strong interferer to baseband. If the envelope of the interferer is constant in time, then a DC offset appears at baseband. In this case, there are several methods known in the art to suppress this unwanted DC offset.
For example, the DC offset may be attenuated by high pass filtering the baseband output of the mixers. While this method is effective to eliminate a DC offset, it is ineffective for distortion due to a non-constant envelope of an interferer. Thus, a need exists for a direct conversion receiver that is capable of attenuating distortion caused by either a constant or a non-constant envelope of an interferer.
Another problem with direct conversion receivers is spurious emissions. The main source of spurious emissions in a direct conversion receiver is local oscillator leakage. In an ordinary superheterodyne receiver, the local oscillator leakage to the antenna is attenuated by the first receiver bandpass filter. In a direct conversion receiver, however, the local oscillator frequency lies within the pass band of the preselector filter. Thus, local oscillator leakage is not suppressed in the traditional direct conversion receiver. Other problems associated with direct conversion receivers are baseband offsets, low frequency noise, and drift.
One approach to address the problems related to those discussed above is set forth in an article entitled “GPS Receiver RF Subsystem Design Overview,” by Hammell et al. The “GPS Receiver RF Subsystem Design Overview” article discloses a receiver which uses a direct-sequence PN “T”-code to modulate the local oscillator. This results in simultaneously frequency conversion and modulation of the input signal to an intermediate frequency. The modulated IF signal is then amplified and bandpass filtered then despread using the same “T”-code to separate the desired signal from the local oscillation leakage signal. This configuration provides a receiver capable of rejecting unwanted LO leakage from the RF to IF in a heterodyne receiver.
Although the configuration of the “GPS Receiver RF Subsystem Design Overview” article reduces out-of-band interference, the disclosed receiver fails to substantially lower DC offsets while maintaining high spurious response rejection.
Another approach to address the problems discussed above is set forth in U.S. Pat. No. 6,192,225 to Arpaia et al. U.S. Pat. No. 6,192,225 discloses a direct conversion receiver which uses a local oscillator to produce a local reference signal at the frequency of the modulation of the input signal. The local oscillation signal is modified by switchable phase elements, and the modified local oscillation signals are then fed to an I-channel mixer and a Q-channel mixer. The oscillation signals going to the two mixers are out of phase with each other due to the modification by the phase change elements. The mixed signals are fed to switchable inverters and low pass filters.
Although the direct conversion receiver of U.S. Pat. No. 6,192,225 reduces spurious emissions, the disclosed direct conversion receiver fails to substantially lower DC offsets or reduce sensitivity to amplitude modulation (AM) while maintaining high spurious response rejection.
A further approach to address the problems discussed above is set forth in U.S. Pat. No. 6,125,272 to Bautista et al. U.S. Pat. No. 6,125,275 discloses a differential RF mixer circuit that employs dynamic matching. The mixer circuit utilizes a switching network that includes cyclically changing transistor pairs to create an imbalance in the circuit that can be modulated. The modulation can be used to eliminate the IM2 distortion element from the desired signal.
Although the differential RF mixer circuit of U.S. Pat. No. 6,125,272 reduces the IM2 distortion element from the desired signal, the disclosed differential RF mixer circuit fails to substantially lower DC offsets or reduce sensitivity to amplitude modulation (AM) while maintaining high spurious response rejection.
A third approach is the use of “chopping” or “dynamic matching”. “Chopping” or “dynamic matching” at two or more ports of a mixer can potentially mitigate sources of baseband offsets, low frequency noise, and drift associated with a direct conversion receiver. One example of a conventional “chopping” or “dynamic matching” approach is illustrated in FIG. 2.
In the “chopped” or “dynamic matched” mixer of FIG. 2, the “chop” 10 and “dechop” 12 blocks multiply the analog waveform by +/−1 under control of the logic level with a frequency of Fchop. In this conventional example, a radio frequency signal, RF-In, is fed to the “chop” block 10 where to signal is modulated with a frequency of Fchop. The modulated signal is then mixed with a local oscillation signal, LO-In, at mixer 11. The mixed signal is de-modulated by “dechop” block 12 with a frequency of Fchop.
The “chopped” or “dynamic matched” mixer of FIG. 2 is easily implemented in a CMOS or BiCMOS process since good, low distortion switches are capable of implementing a +/−1 multiplication with good result in a fully-differential implementation.
The primarily penalty associated with “chopping” or “dynamic matching” is susceptibility to spurious responses at FLO+/−n*FCHOP. In an ideal implementation, these responses are all zero. However, a real circuit is nonlinear during the transitions, which are finite in duration, and thus the responses are not all non-zero, and result in spurious responses.
Furthermore, in a direct conversion receiver, the local oscillator produces a DC output due to self-mixing in the RF port. The offset changes due to the dynamic loading of the antenna by the environment as well as changes in temperature, supply voltage, and/or channel number (local oscillator frequency). Most of the change is at very low frequencies and can be removed easily using a feedback loop.
However, only slow changes in the DC can be removed by the feedback loop, not higher speed changes in the DC offset since these higher speed changes fall outside the bandwidth of the tracking loop.
In the situation where global systems for mobile communication technology is being utilized, the gain from the antenna to the mixer is typically 10 db. The required sensitivity for the receiver is typically −106 dBm (1.1 microvolts-rms). The rms signal voltage at the mixer is 3.5 microvolts. In order not to cause reduced sensitivity, the DC offset due to the mixer needs to be about 1.1 microvolts or less. However measurements show that typically, the offset when referenced to the mixer input is 1 to 2 mVrms. Ideally, 60 dB of DC offset suppression is needed.
Moreover, a regenerative divider can be used to synthesize the local oscillator and/or an even-harmonic mixer can be used so that the common direct conversion receiver effects of local oscillator pulling and AM rectification can be minimized. However, radiation and dynamic re-reflection of the local oscillator signal at the antenna is still a significant problem.
Therefore, it is desirable to provide a direct conversion receiver that avoids the various problems outlined above. Moreover, it is desirable to provide a direct conversion receiver that spreads an intermediate frequency (IF) across a wide bandwidth and despreads the IF using the same modulated signal to lower DC offsets and to reduce sensitivity to amplitude modulation (AM), while maintaining high spurious response rejection.